Method and device for adaptive bandwidth pitch search in coding wideband signals

ABSTRACT

A pitch search method and device for digitally encoding a wideband signal, in particular but not exclusively a speech signal, in view of transmitting, or storing, and synthesizing this wideband sound signal. The new method and device which achieve efficient modeling of the harmonic structure of the speech spectrum uses several forms of low pass filters applied to a pitch codevector, the one yielding higher prediction gain (i.e. the lowest pitch prediction error) is selected and the associated pitch codebook parameters are forwarded.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an efficient technique for digitallyencoding a wideband signal, in particular but not exclusively a speechsignal, in view of transmitting, or storing, and synthesizing thiswideband sound signal. More specifically, this invention deals with animproved pitch search device and method.

2. Brief description of the prior art:

The demand for efficient digital wideband speech/audio encodingtechniques with a good subjective quality/bit rate trade-off isincreasing for numerous applications such as audio/videoteleconferencing, multimedia, and wireless applications, as well asInternet and packet network applications. Until recently, telephonebandwidths filtered in the range 200-3400 Hz were mainly used in speechcoding applications. However, there is an increasing demand for widebandspeech applications in order to increase the intelligibility andnaturalness of the speech signals. A bandwidth in the range 50-7000 Hzwas found sufficient for delivering a face-to-face speech quality. Foraudio signals, this range gives an acceptable audio quality, but stilllower than the CD quality which operates on the range 20-20000 Hz.

A speech encoder converts a speech signal into a digital bitstream whichis transmitted over a communication channel (or stored in a storagemedium). The speech signal is digitized (sampled and quantized withusually 16-bits per sample) and the speech encoder has the role ofrepresenting these digital samples with a smaller number of bits whilemaintaining a good subjective speech quality. The speech decoder orsynthesizer operates on the transmitted or stored bit stream andconverts it back to a sound signal.

One of the best prior art techniques capable of achieving a goodquality/bit rate trade-off is the so-called Code Excited LinearPrediction (CELP) technique. According to this technique, the sampledspeech signal is processed in successive blocks of L samples usuallycalled frames where L is some predetermined number (corresponding to10-30 ms of speech). In CELP, a linear prediction (LP) filter iscomputed and transmitted every frame. The L-sample frame is then dividedinto smaller blocks called subframes of size N samples, where L=kN and kis the number of subframes in a frame (N usually corresponds to 4-10 msof speech). An excitation signal is determined in each subframe, whichusually consists of two components: one from the past excitation (alsocalled pitch contribution or adaptive codebook) and the other from aninnovation codebook (also called fixed codebook). This excitation signalis transmitted and used at the decoder as the input of the LP synthesisfilter in order to obtain the synthesized speech.

An innovation codebook in the CELP context, is an indexed set ofN-sample-long sequences which will be referred to as N-dimensionalcodevectors. Each codebook sequence is indexed by an integer k rangingfrom 1 to M where M represents the size of the codebook often expressedas a number of bits b, where M=2^(b).

To synthesize speech according to the CELP technique, each block of Nsamples is synthesized by filtering an appropriate codevector from acodebook through time varying filters modeling the spectralcharacteristics of the speech signal. At the encoder end, the syntheticoutput is computed for all, or a subset, of the codevectors from thecodebook (codebook search). The retained codevector is the one producingthe synthetic output closest to the original speech signal according toa perceptually weighted distortion measure. This perceptual weighting isperformed using a so-called perceptual weighting filter, which isusually derived from the LP filter.

The CELP model has been very successful in encoding telephone band soundsignals, and several CELP-based standards exist in a wide range ofapplications, especially in digital cellular applications. In thetelephone band, the sound signal is band-limited to 200-3400 Hz andsampled at 8000 samples/sec. In wideband speech/audio applications, thesound signal is band-limited to 50-7000 Hz and sampled at 16000samples/sec.

Some difficulties arise when applying the telephone-band optimized CELPmodel to wideband signals, and additional features need to be added tothe model in order to obtain high quality wideband signals. Widebandsignals exhibit a much wider dynamic range compared to telephone-bandsignals, which results in precision problems when a fixed-pointimplementation of the algorithm is required (which is essential inwireless applications). Further, the CELP model will often spend most ofits encoding bits on the low-frequency region, which usually has higherenergy contents, resulting in a low-pass output signal. To overcome thisproblem, the perceptual weighting filter has to be modified in order tosuit wideband signals, and pre-emphasis techniques which boost the highfrequency regions become important to reduce the dynamic range, yieldinga simpler fixed-point implementation, and to ensure a better encoding ofthe higher frequency contents of the signal. Further, the pitch contentsin the spectrum of voiced segments in wideband signals do not extendover the whole spectrum range, and the amount of voicing shows morevariation compared to narrow-band signals. Therefore, in case ofwideband signals, existing pitch search structures are not veryefficient. Thus, it is important to improve the closed-loop pitchanalysis to better accommodate the variations in the voicing level.

OBJECTS OF THE INVENTION

An object of the present invention is therefore to provide a method anddevice for efficiently encoding wideband (7000 Hz) sound signals usingCELP-type encoding techniques, using improved pitch analysis in order toobtain high a quality reconstructed sound signal.

SUMMARY OF THE INVENTION

More specifically, in accordance with the present invention, there isprovided a method for selecting an optimal set of pitch codebookparameters associated to a signal path, from at least two signal paths,having the lowest calculated pitch prediction error. The pitchprediction error is calculated in response to a pitch codevector from apitch codebook search device. In at least one of the two signal paths,the pitch prediction error is filtered before supplying the pitchcodevector for calculation of said pitch prediction error of said onepath. Finally, the pitch prediction errors calculated in said at leasttwo signal paths are compared, the signal path having the lowestcalculated pitch prediction error is chosen, and the set of pitchcodebook parameters associated to the choosen signal path are selected.

The pitch analysis device of the invention, for producing an optimal setof pitch codebook parameters, comprises:

-   -   a) at least two signal paths associated to respective sets of        pitch codebook parameters, wherein:        -   i) each signal path comprises a pitch prediction error            calculating device for calculating a pitch prediction error            of a pitch codevector from a pitch codebook search device;            and        -   ii) at least one of the two paths comprises a filter for            filtering the pitch codevector before supplying the pitch            codevector to the path's pitch prediction error calculating            device; and    -   b) a selector for comparing the pitch prediction errors        calculated in the signal paths, for choosing the signal path        having the lowest calculated pitch prediction error, and for        selecting the set of pitch codebook parameters associated to the        choosen signal path.

The new method and device which achieve efficient modeling of theharmonic structure of the speech spectrum uses several forms of low passfilters applied to the past excitation and the one yielding higherprediction gain is selected. When subsample pitch resolution is used,the low pass filters can be incorporated into the interpolation filtersused to obtain the higher pitch resolution.

In a preferred embodiment of the invention, each pitch prediction errorcalculating device of the pitch analysis device described abovecomprises:

-   -   a) a convolution unit for convolving the pitch codevector with a        weighted synthesis filter impulse response signal and therefore        calculating a convolved pitch codevector;    -   b) a pitch gain calculator for calculating a pitch gain in        response to the convolved pitch codevector and a pitch search        target vector;    -   c) an amplifier for multiplying the convolved pitch codevector        by the pitch gain to thereby produce an amplified convolved        pitch codevector; and    -   d) a combiner circuit for combining the amplified convolved        pitch codevector with the pitch search target vector to thereby        produce the pitch prediction error.

In another preferred embodiment of the invention, the pitch gaincalculator comprises a means for calculating said pitch gain b^((j))using the relation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths,and where x is said pitch search target vector, and y^((j)) is saidconvolved pitch codevector.

The present invnetion further relates to an encoder, having the pitchanalysis device described above, for encoding a wideband input signaland comprising:

-   -   a) a linear prediction synthesis filter calculator responsive to        the wideband signal for producing linear prediction synthesis        filter coefficients;    -   b) a perceptual weighting filter, responsive to the wideband        signal and the linear prediction synthesis filter coefficients,        for producing a perceptually weighted signal;    -   c) an impulse response generator responsive to the linear        prediction synthesis filter coefficients for producing a        weighted synthesis filter impulse response signal;    -   d) a pitch search unit for producing pitch codebook parameters,        comprising:        -   i) a pitch codebook search device responsive to the            perceptually weighted signal and the linear prediction            synthesis filter coefficients for producing the pitch            codevector and an innovative search target vector; and        -   ii) the pitch analysis device responsive to the pitch            codevector for selecting, from the sets of pitch codebook            parameters, the set of pitch codebook parameters associated            to the path having the lowest calculated pitch prediction            error;    -   d) an innovative codebook search device, responsive to the        weighted synthesis filter impulse response signal, and the        innovative search target vector, for producing innovative        codebook parameters; and    -   e) a signal forming device for producing an encoded wideband        signal comprising the set of pitch codebook parameters        associated to the path having the lowest pitch prediction error,        the innovative codebook parameters, and the linear prediction        synthesis filter coefficients.

The present invention still further relates to a cellular communicationsystem, a cellular mobile transmitter/receiver unit, a cellular networkelement, and a bidirectional wireless communication sub-systemcomprising the above described decoder.

The objects, advantages and other features of the present invention willbecome more apparent upon reading of the following non restrictivedescription of a preferred embodiment thereof, given by way of exampleonly with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

In the appended drawings:

FIG. 1 is a schematic block diagram of a preferred embodiment ofwideband encoding device;

FIG. 2 is a schematic block diagram of a preferred embodiment ofwideband decoding device;

FIG. 3 is a schematic block diagram of a preferred embodiment of pitchanalysis device; and

FIG. 4 is a simplified, schematic block diagram of a cellularcommunication system in which the wideband encoding device of FIG. 1 andthe wideband decoding device of FIG. 2 can be used.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

As well known to those of ordinary skill in the art, a cellularcommunication system such as 401 (see FIG. 4) provides atelecommunication service over a large geographic area by dividing thatlarge geographic area into a number C of smaller cells. The C smallercells are serviced by respective cellular base stations 402 ₁, 402 ₂ . .. 402 _(c) to provide each cell with radio signalling, audio and datachannels.

Radio signalling channels are used to page mobile radiotelephones(mobile transmitter/receiver units) such as 403 within the limits of thecoverage area (cell) of the cellular base station 402, and to placecalls to other radiotelephones 403 located either inside or outside thebase station's cell or to another network such as the Public SwitchedTelephone Network (PSTN) 404.

Once a radiotelephone 403 has successfully placed or received a call, anaudio or data channel is established between this radiotelephone 403 andthe cellular base station 402 corresponding to the cell in which theradiotelephone 403 is situated, and communication between the basestation 402 and radiotelephone 403 is conducted over that audio or datachannel. The radiotelephone 403 may also receive control or timinginformation over a signalling channel while a call is in progress.

If a radiotelephone 403 leaves a cell and enters another adjacent cellwhile a call is in progress, the radiotelephone 403 hands over the callto an available audio or data channel of the new cell base station 402.If a radiotelephone 403 leaves a cell and enters another adjacent cellwhile no call is in progress, the radiotelephone 403 sends a controlmessage over the signalling channel to log into the base station 402 ofthe new cell. In this manner mobile communication over a widegeographical area is possible.

The cellular communication system 401 further comprises a controlterminal 405 to control communication between the cellular base stations402 and the PSTN 404, for example during a communication between aradiotelephone 403 and the PSTN 404, or between a radiotelephone 403located in a first cell and a radiotelephone 403 situated in a secondcell.

Of course, a bidirectional wireless radio communication subsystem isrequired to establish an audio or data channel between a base station402 of one cell and a radiotelephone 403 located in that cell. Asillustrated in very simplified form in FIG. 4, such a bidirectionalwireless radio communication subsystem typically comprises in theradiotelephone 403:

-   -   a transmitter 406 including:        -   an encoder 407 for encoding the voice signal; and        -   a transmission circuit 408 for transmitting the encoded            voice signal from the encoder 407 through an antenna such as            409; and    -   a receiver 410 including:        -   a receiving circuit 411 for receiving a transmitted encoded            voice signal usually through the same antenna 409; and        -   a decoder 412 for decoding the received encoded voice signal            from the receiving circuit 411.

The radiotelephone further comprises other conventional radiotelephonecircuits 413 to which the encoder 407 and decoder 412 are connected andfor processing signals therefrom, which circuits 413 are well known tothose of ordinary skill in the art and, accordingly, will not be furtherdescribed in the present specification.

Also, such a bidirectional wireless radio communication subsystemtypically comprises in the base station 402:

-   -   a transmitter 414 including:        -   an encoder 415 for encoding the voice signal; and        -   a transmission circuit 416 for transmitting the encoded            voice signal from the encoder 415 through an antenna such as            417; and    -   a receiver 418 including:        -   a receiving circuit 419 for receiving a transmitted encoded            voice signal through the same antenna 417 or through another            antenna (not shown); and        -   a decoder 420 for decoding the received encoded voice signal            from the receiving circuit 419.

The base station 402 further comprises, typically, a base stationcontroller 421, along with its associated database 422, for controllingcommunication between the control terminal 405 and the transmitter 414and receiver 418.

As well known to those of ordinary skill in the art, voice encoding isrequired in order to reduce the bandwidth necessary to transmit soundsignal, for example voice signal such as speech, across thebidirectional wireless radio communication subsystem, i.e., between aradiotelephone 403 and a base station 402.

LP voice encoders (such as 415 and 407) typically operating at 13kbits/second and below such as Code-Excited Linear Prediction (CELP)encoders typically use a LP synthesis filter to model the short-termspectral envelope of the voice signal. The LP information istransmitted, typically, every 10 or 20 ms to the decoder (such 420 and412) and is extracted at the decoder end.

The novel techniques disclosed in the present specification may apply todifferent LP-based coding systems. However, a CELP-type coding system isused in the preferred embodiment for the purpose of presenting anon-limitative illustration of these techniques. In the same manner,such techniques can be used with sound signals other than voice andspeech as well with other types of wideband signals.

FIG. 1 shows a general block diagram of a CELP-type speech encodingdevice 100 modified to better accommodate wideband signals.

The sampled input speech signal 114 is divided into successive L-sampleblocks called “frames”. In each frame, different parameters representingthe speech signal in the frame are computed, encoded, and transmitted.LP parameters representing the LP synthesis filter are usually computedonce every frame. The frame is further divided into smaller blocks of Nsamples (blocks of length N), in which excitation parameters (pitch andinnovation) are determined. In the CELP literature, these blocks oflength N are called “subframes” and the N-sample signals in thesubframes are referred to as N-dimensional vectors. In this preferredembodiment, the length N corresponds to 5 ms while the length Lcorresponds to 20 ms, which means that a frame contains four subframes(N=80 at the sampling rate of 16 kHz and 64 after down-sampling to 12.8kHz). Various N-dimensional vectors occur in the encoding procedure. Alist of the vectors which appear in FIGS. 1 and 2 as well as a list oftransmitted parameters are given herein below:

List of the Main N-Dimensional Vectors

-   -   s Wideband signal input speech vector (after down-sampling,        pre-processing, and preemphasis);    -   s_(w) Weighted speech vector;    -   s₀ Zero-input response of weighted synthesis filter,    -   s_(p) Down-sampled pre-processed signal;    -   Oversampled synthesized speech signal;    -   s′ Synthesis signal before deemphasis;    -   S_(d) Deemphasized synthesis signal;    -   s_(h) Synthesis signal after deemphasis and postprocessing;    -   x Target vector for pitch search;    -   x′ Target vector for innovation search;    -   h Weighted synthesis filter impulse response;    -   v_(T) Adaptive (pitch) codebook vector at delay T;    -   y_(T) Filtered pitch codebook vector (v_(T) convolved with h);    -   c_(k) Innovative codevector at index k (k-th entry from the        innovation codebook);    -   c_(f) Enhanced scaled innovation codevector;    -   u Excitation signal (scaled innovation and pitch codevectors);    -   u′ Enhanced excitation;    -   z Band-pass noise sequence;    -   w′ White noise sequence; and    -   w Scaled noise sequence.

List of Transmitted Parameters

-   -   STP Short term prediction parameters (defining A(z));    -   T Pitch lag (or pitch codebook index);    -   b Pitch gain (or pitch codebook gain);    -   j Index of the low-pass filter used on the pitch codevector;    -   k Codevector index (innovation codebook entry); and    -   g Innovation codebook gain.

In this preferred embodiment, the STP parameters are transmitted onceper frame and the rest of the parameters are transmitted four times perframe (every subframe).

Encoder Side

The sampled speech signal is encoded on a block by block basis by theencoding device 100 of FIG. 1 which is broken down into eleven modulesnumbered from 101 to 111.

The input speech is processed into the above mentioned L-sample blockscalled frames.

Referring to FIG. 1, the sampled input speech signal 114 is down-sampledin a down-sampling module 101. For example, the signal is down-sampledfrom 16 kHz down to 12.8 kHz, using techniques well known to those ofordinary skill in the art. Down-sampling down to another frequency canof course be envisaged. Down-sampling increases the coding efficiency,since a smaller frequency bandwidth is encoded. This also reduces thealgorithmic complexity since the number of samples in a frame isdecreased. The use of down-sampling becomes significant when the bitrate is reduced below 16 kbit/s, although down-sampling is not essentialabove 16 kbit/s.

After down-sampling, the 320-sample frame of 20 ms is reduced to256-sample frame (down-sampling ratio of ⅘).

The input frame is then supplied to the optional pre-processing block102. Pre-processing block 102 may consist of a high-pass filter with a50 Hz cut-off frequency. High-pass filter 102 removes the unwanted soundcomponents below 50 Hz.

The down-sampled pre-processed signal is denoted by s_(p)(n), n=0, 1, 2,. . . , L−1, where L is the length of the frame (256 at a samplingfrequency of 12.8 kHz). In a preferred embodiment of the preemphasisfilter 103, the signal s_(p)(n) is preemphasized using a filter havingthe following transfer function:P(z)=1−μz ⁻¹where μ is a preemphasis factor with a value located between 0 and 1 (atypical value is μ=0.7). A higher-order filter could also be used. Itshould be pointed out that high-pass filter 102 and preemphasis filter103 can be interchanged to obtain more efficient fixed-pointimplementations.

The function of the preemphasis filter 103 is to enhance the highfrequency contents of the input signal. It also reduces the dynamicrange of the input speech signal, which renders it more suitable forfixed-point implementation. Without preemphasis, LP analysis infixed-point using single-precision arithmetic is difficult to implement.

Preemphasis also plays an important role in achieving a proper overallperceptual weighting of the quantization error, which contributes toimproved sound quality. This will be explained in more detail hereinbelow.

The output of the preemphasis filter 103 is denoted s(n). This signal isused for performing LP analysis in calculator module 104. LP analysis isa technique well known to those of ordinary skill in the art. In thispreferred embodiment, the autocorrelation approach is used. In theautocorrelation approach, the signal s(n) is first windowed using aHamming window (having usually a length of the order of 30-40 ms). Theautocorrelations are computed from the windowed signal, andLevinson-Durbin recursion is used to compute LP filter coefficients,a_(i), where i=1, . . . p, and where p is the LP order, which istypically 16 in wideband coding. The parameters a_(i) are thecoefficients of the transfer function of the LP filter, which is givenby the following relation:${A(z)} = {1 + {\sum\limits_{i = 1}^{p}{a_{i}z^{- 1}}}}$

LP analysis is performed in calculator module 104, which also performsthe quantization and interpolation of the LP filter coefficients. The LPfilter coefficients are first transformed into another equivalent domainmore suitable for quantization and interpolation purposes. The linespectral pair (LSP) and immitance spectral pair (ISP) domains are twodomains in which quantization and interpolation can be efficientlyperformed. The 16 LP filter coefficients, a_(i), can be quantized in theorder of 30 to 50 bits using split or multi-stage quantization, or acombination thereof. The purpose of the interpolation is to enableupdating the LP filter coefficients every subframe while transmittingthem once every frame, which improves the encoder performance withoutincreasing the bit rate. Quantization and interpolation of the LP filtercoefficients is believed to be otherwise well known to those of ordinaryskill in the art and, accordingly, will not be further described in thepresent specification.

The following paragraphs will describe the rest of the coding operationsperformed on a subframe basis. In the following description, the filterA(z) denotes the unquantized interpolated LP filter of the subframe, andthe filter Â(z) denotes the quantized interpolated LP filter of thesubframe.

Perceptual Weighting:

In analysis-by-synthesis encoders, the optimum pitch and innovationparameters are searched by minimizing the mean squared error between theinput speech and synthesized speech in a perceptually weighted domain.This is equivalent to minimizing the error between the weighted inputspeech and weighted synthesis speech.

The weighted signal s_(w)(n) is computed in a perceptual weightingfilter 105. Traditionally, the weighted signal s_(w)(n) is computed by aweighting filter having a transfer function W(z) in the form:W(z)=A(z/γ ₁)/A(z/γ ₂) where 0<γ₂<γ₁≦1As well known to those of ordinary skill in the art, in prior artanalysis-by-synthesis (AbS) encoders, analysis shows that thequantization error is weighted by a transfer function W⁻¹(z), which isthe inverse of the transfer function of the perceptual weighting filter105. This result is well described by B. S. Atal and M. R. Schroeder in“Predictive coding of speech and subjective error criteria”, IEEETransaction ASSP, vol. 27, no. 3, pp. 247-254, June 1979. Transferfunction W⁻¹(z) exhibits some of the formant structure of the inputspeech signal. Thus, the masking property of the human ear is exploitedby shaping the quantization error so that it has more energy in theformant regions where it will be masked by the strong signal energypresent in these regions. The amount of weighting is controlled by thefactors γ₁ and γ₂.

The above traditional perceptual weighting filter 105 works well withtelephone band signals. However, it was found that this traditionalperceptual weighting filter 105 is not suitable for efficient perceptualweighting of wideband signals. It was also found that the traditionalperceptual weighting filter 105 has inherent limitations in modellingthe formant structure and the required spectral tilt concurrently. Thespectral tilt is more pronounced in wideband signals due to the widedynamic range between low and high frequencies. The prior art hassuggested to add a tilt filter into W(z) in order to control the tiltand formant weighting of the wideband input signal separately.

A novel solution to this problem is, in accordance with the presentinvention, to introduce the preemphasis filter 103 at the input, computethe LP filter A(z) based on the preemphasized speech s(n), and use amodified filter W(z) by fixing its denominator.

LP analysis is performed in module 104 on the preemphasized signal s(n)to obtain the LP filter A(z). Also, a new perceptual weighting filter105 with fixed denominator is used. An example of transfer function forthe perceptual weighting filter 104 is given by the following relation:W(z)=A(z/γ ₁)/(1−γ₂ z ⁻¹) where 0<γ₂<γ₁≦1A higher order can be used at the denominator. This structuresubstantially decouples the formant weighting from the tilt.

Note that because A(z) is computed based on the preemphasized speechsignal s(n), the tilt of the filter 1/A(z/γ₁) is less pronouncedcompared to the case when A(z) is computed based on the original speech.Since deemphasis is performed at the decoder end using a filter havingthe transfer function:P ⁻¹(z)=1/(1−μz ⁻¹),the quantization error spectrum is shaped by a filter having a transferfunction W⁻¹(z)P⁻¹(z). When γ₂ is set equal to μ, which is typically thecase, the spectrum of the quantization error is shaped by a filter whosetransfer function is 1/A(z/γ₁), with A(z) computed based on thepreemphasized speech signal. Subjective listening showed that thisstructure for achieving the error shaping by a combination ofpreemphasis and modified weighting filtering is very efficient forencoding wideband signals, in addition to the advantages of ease offixed-point algorithmic implementation.Pitch Analysis:

In order to simplify the pitch analysis, an open-loop pitch lag T_(OL)is first estimated in the open-loop pitch search module 106 using theweighted speech signal s_(w)(n). Then the closed-loop pitch analysis,which is performed in closed-loop pitch search module 107 on a subframebasis, is restricted around the open-loop pitch lag T_(OL) whichsignificantly reduces the search complexity of the LTP parameters T andb (pitch lag and pitch gain). Open-loop pitch analysis is usuallyperformed in module 106 once every 10 ms (two subframes) usingtechniques well known to those of ordinary skill in the art.

The target vector x for LTP (Long Term Prediction) analysis is firstcomputed. This is usually done by subtracting the zero-input response s₀of weighted synthesis filter W(z)/Â(z) from the weighted speech signals_(w)(n). This zero-input response s₀ is calculated by a zero-inputresponse calculator 108. More specifically, the target vector x iscalculated using the following relation:x=s _(w) −s ₀where x is the N-dimensional target vector, s_(w) is the weighted speechvector in the subframe, and so is the zero-input response of filterW(z)/Â(z) which is the output of the combined filter W(z)/Â(z) due toits initial states. The zero-input response calculator 108 is responsiveto the quantized interpolated LP filter Â(z) from the LP analysis,quantization and interpolation calculator 104 and to the initial statesof the weighted synthesis filter W(z)/Â(z) stored in memory module 111to calculate the zero-input response s₀ (that part of the response dueto the initial states as determined by setting the inputs equal to zero)of filter W(z)/Â(z). This operation is well known to those of ordinaryskill in the art and, accordingly, will not be further described.

Of course, alternative but mathematically equivalent approaches can beused to compute the target vector x.

A N-dimensional impulse response vector h of the weighted synthesisfilter W(z)/Â(z) is computed in the impulse response generator 109 usingthe LP filter coefficients A(z) and Â(z) from module 104. Again, thisoperation is well known to those of ordinary skill in the art and,accordingly, will not be further described in the present specification.

The closed-loop pitch (or pitch codebook) parameters b, T and j arecomputed in the closed-loop pitch search module 107, which uses thetarget vector x, the impulse response vector h and the open-loop pitchlag T_(OL) as inputs. Traditionally, the pitch prediction has beenrepresented by a pitch filter having the following transfer function:1/(1−bz^(−T))where b is the pitch gain and T is the pitch delay or lag. In this case,the pitch contribution to the excitation signal u(n) is given bybu(n−T), where the total excitation is given byu(n)=bu(n−T)+gc _(k)(n)with g being the innovative codebook gain and c_(k)(n) the innovativecodevector at index k.

This representation has limitations if the pitch lag T is shorter thanthe subframe length N. In another representation, the pitch contributioncan be seen as a pitch codebook containing the past excitation signal.Generally, each vector in the pitch codebook is a shift-by-one versionof the previous vector (discarding one sample and adding a new sample).For pitch lags T>N, the pitch codebook is equivalent to the filterstructure (1/(1−bz^(−T)), and a pitch codebook vector v_(T)(n) at pitchlag T is given byV _(T)(n)=u(n−T), n=0, . . . , N−1.For pitch lags T shorter than N, a vector v_(T)(n) is built by repeatingthe available samples from the past excitation until the vector iscompleted (this is not equivalent to the filter structure).

In recent encoders, a higher pitch resolution is used whichsignificantly improves the quality of voiced sound segments. This isachieved by oversampling the past excitation signal using polyphaseinterpolation filters. In this case, the vector v_(T)(n) usuallycorresponds to an interpolated version of the past excitation, withpitch lag T being a non-integer delay (e.g. 50.25).

The pitch search consists of finding the best pitch lag T and gain bthat minimize the mean squared weighted error E between the targetvector x and the scaled filtered past excitation. Error E beingexpressed as:E=∥x−by _(T)∥²where y_(T) is the filtered pitch codebook vector at pitch lag T:${{y_{T}(n)} = {{{v_{T}(n)}*{h(n)}} = {\sum\limits_{i = o}^{n}{{v_{T}(i)}{h\left( {n - i} \right)}}}}},\quad{n = 0},\ldots\quad,{N - 1.}$

It can be shown that the error E is minimized by maximizing the searchcriterion $C = \frac{x^{t}y_{T}}{\sqrt{y_{T}^{t}y_{T}}}$where t denotes vector transpose.

In the preferred embodiment of the present invention, a ⅓ subsamplepitch resolution is used, and the pitch (pitch codebook) search iscomposed of three stages.

In the first stage, an open-loop pitch lag T_(OL) is estimated inopen-loop pitch search module 106 in response to the weighted speechsignal s_(w)(n). As indicated in the foregoing description, thisopen-loop pitch analysis is usually performed once every 10 ms (twosubframes) using techniques well known to those of ordinary skill in theart.

In the second stage, the search criterion C is searched in theclosed-loop pitch search module 107 for integer pitch lags around theestimated open-loop pitch lag T_(OL) (usually ±5), which significantlysimplifies the search procedure. A simple procedure is used for updatingthe filtered codevector YT without the need to compute the convolutionfor every pitch lag.

Once an optimum integer pitch lag is found in the second stage, a thirdstage of the search (module 107) tests the fractions around that optimuminteger pitch lag.

When the pitch predictor is represented by a filter of the form1/(1−bz^(−T)), which is a valid assumption for pitch lags T>N, thespectrum of the pitch filter exhibits a harmonic structure over theentire frequency range, with a harmonic frequency related to 1/T. Incase of wideband signals, this structure is not very efficient since theharmonic structure in wideband signals does not cover the entireextended spectrum. The harmonic structure exists only up to a certainfrequency, depending on the speech segment. Thus, in order to achieveefficient representation of the pitch contribution in voiced segments ofwideband speech, the pitch prediction filter needs to have theflexibility of varying the amount of periodicity over the widebandspectrum.

A new method which achieves efficient modeling of the harmonic structureof the speech spectrum of wideband signals is disclosed in the presentspecification, whereby several forms of low pass filters are applied tothe past excitation and the low pass filter with higher prediction gainis selected.

When subsample pitch resolution is used, the low pass filters can beincorporated into the interpolation filters used to obtain the higherpitch resolution. In this case, the third stage of the pitch search, inwhich the fractions around the chosen integer pitch lag are tested, isrepeated for the several interpolation filters having different low-passcharacteristics and the fraction and filter index which maximize thesearch criterion C are selected.

A simpler approach is to complete the search in the three stagesdescribed above to determine the optimum fractional pitch lag using onlyone interpolation filter with a certain frequency response, and selectthe optimum low-pass filter shape at the end by applying the differentpredetermined low-pass filters to the chosen pitch codebook vector v_(T)and select the low-pass filter which minimizes the pitch predictionerror. This approach is discussed in detail below.

FIG. 3 illustrates a schematic block diagram of a preferred embodimentof the proposed approach.

In memory module 303, the past excitation signal u(n), n<0, is stored.The pitch codebook search module 301 is responsive to the target vectorx, to the open-loop pitch lag TOL and to the past excitation signalu(n), n<0, from memory module 303 to conduct a pitch codebook (pitchcodebook) search minimizing the above-defined search criterion C. Fromthe result of the search conducted in module 301, module 302 generatesthe optimum pitch codebook vector v_(T). Note that since a sub-samplepitch resolution is used (fractional pitch), the past excitation signalu(n), n<0, is interpolated and the pitch codebook vector v_(T)corresponds to the interpolated past excitation signal. In thispreferred embodiment, the interpolation filter (in module 301, but notshown) has a low-pass filter characteristic removing the frequencycontents above 7000 Hz.

In a preferred embodiment, K filter characteristics are used; thesefilter characteristics could be low-pass or band-pass filtercharacteristics. Once the optimum codevector v_(T) is determined andsupplied by the pitch codevector generator 302, K filtered versions ofv_(T) are computed respectively using K different frequency shapingfilters such as 305 ^((j)), where j=1, 2, . . . , K. These filteredversions are denoted v_(f) ^((j)), where j=1, 2, . . . , K. Thedifferent vectors v_(f) ^((j)) are convolved in respective modules 304^((j)), where j=0, 1, 2, . . . , K, with the impulse response h toobtain the vectors y^((j)), where j=0, 1, 2, . . . , K. To calculate themean squared pitch prediction error for each vector y^((j)), the valuey^((j)) is multiplied by the gain b by means of a correspondingamplifier 307 ^((j)) and the value by^((j)) is subtracted from thetarget vector x by means of a corresponding subtractor 308 ^((j)).Selector 309 selects the frequency shaping filter 305 ^((j)) whichminimizes the mean squared pitch prediction errore ^((j)) =∥x−b ^((j)) y ^((j))∥², j=1, 2, . . . , KTo calculate the mean squared pitch prediction error e^((j)) for eachvalue of y^((j)), the value y^((j)) is multiplied by the gain b by meansof a corresponding amplifier 307 ^((j)) and the value b^((j))y^((j)) issubtracted from the target vector x by means of subtractors 308 ^((j)).Each gain b^((j)) is calculated in a corresponging gain calculator 306^((j)) in association with the frequency shaping filter at index j,using the following relationship:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥².

In selector 309, the parameters b, T, and j are chosen based on v_(T) orv_(f) ^((j)) which minimizes the mean squared pitch prediction error e.

Referring back to FIG. 1, the pitch codebook index T is encoded andtransmitted to multiplexer 112. The pitch gain b is quantized andtransmitted to multiplexer 112. With this new approach, extrainformation is needed to encode the index j of the selected frequencyshaping filter in multiplexer 112. For example, if three filters areused (j=0, 1, 2, 3), then two bits are needed to represent thisinformation. The filter index information j can also be encoded jointlywith the pitch gain b.

Innovative Codebook Search:

Once the pitch, or LTP (Long Term Prediction) parameters b, T, and j aredetermined, the next step is to search for the optimum innovativeexcitation by means of search module 110 of FIG. 1. First, the targetvector x is updated by subtracting the LTP contribution:x′=x−by _(T)where b is the pitch gain and y_(T) is the filtered pitch codebookvector (the past excitation at delay T filtered with the selected lowpass filter and convolved with the inpulse response h as described withreference to FIG. 3).

The search procedure in CELP is performed by finding the optimumexcitation codevector c_(k) and gain g which minimize the mean-squarederror between the target vector and the scaled filtered codevectorE=∥x′−gHc _(k)∥²where H is a lower triangular convolution matrix derived from theimpulse response vector h.

In the preferred embodiment of the present invention, the innovativecodebook search is performed in module 110 by means of an algebraiccodebook as described in U.S. Pat. Nos. 5,444,816 (Adoul et al.) issuedon Aug. 22, 1995; 5,699,482 granted to Adoul et al., on Dec. 17, 1997;5,754,976 granted to Adoul et al., on May 19, 1998; and 5,701,392 (Adoulet al.) dated Dec. 23, 1997.

Once the optimum excitation codevector c_(k) and its gain g are chosenby module 110, the codebook index k and gain g are encoded andtransmitted to multiplexer 112.

Referring to FIG. 1, the parameters b, T, j, Â(z), k and g aremultiplexed through the multiplexer 112 before being transmitted througha communication channel.

Memory Update:

In memory module 111 (FIG. 1), the states of the weighted synthesisfilter W(z)/Â(z) are updated by filtering the excitation signalu=gc_(k)+bv_(T) through the weighted synthesis filter. After thisfiltering, the states of the filter are memorized and used in the nextsubframe as initial states for computing the zero-input response incalculator module 108.

As in the case of the target vector x, other alternative butmathematically equivalent approaches well known to those of ordinaryskill in the art can be used to update the filter states.

Decoder Side

The speech decoding device 200 of FIG. 2 illustrates the various stepscarried out between the digital input 222 (input stream to thedemultiplexer 217) and the output sampled speech 223 (output of theadder 221).

Demultiplexer 217 extracts the synthesis model parameters from thebinary information received from a digital input channel. From eachreceived binary frame, the extracted parameters are:

-   -   the short-term prediction parameters (STP) Â(z) (once per        frame);    -   the long-term prediction (LTP) parameters T, b, and j (for each        subframe); and    -   the innovation codebook index k and gain g (for each subframe).        The current speech signal is synthesized based on these        parameters as will be explained hereinbelow.

The innovative codebook 218 is responsive to the index k to produce theinnovation codevector c^(k), which is scaled by the decoded gain factorg through an amplifier 224. In the preferred embodiment, an innovativecodebook 218 as described in the above mentioned U.S. Pat. Nos.5,444,816; 5,699,482; 5,754,976; and 5,701,392 is used to represent theinnovative codevector c_(k).

The generated scaled codevector gc_(k) at the output of the amplifier224 is processed through a innovation filter 205.

Periodicity Enhancement:

The generated scaled codevector at the output of the amplifier 224 isprocessed through a frequency-dependent pitch enhancer 205.

Enhancing the periodicity of the excitation signal u improves thequality in case of voiced segments. This was done in the past byfiltering the innovation vector from the innovative codebook (fixedcodebook) 218 through a filter in the form 1/(1−εbz^(−T)) where ε is afactor below 0.5 which controls the amount of introduced periodicity.This approach is less efficient in case of wideband signals since itintroduces periodicity over the entire spectrum. A new alternativeapproach, which is part of the present invention, is disclosed wherebyperiodicity enhancement is achieved by filtering the innovativecodevector c_(k) from the innovative (fixed) codebook through aninnovation filter 205 (F(z)) whose frequency response emphasizes thehigher frequencies more than lower frequencies. The coefficients of F(z)are related to the amount of periodicity in the excitation signal u.

Many methods known to those skilled in the art are available forobtaining valid periodicity coefficients. For example, the value of gainb provides an indication of periodicity. That is, if gain b is close to1, the periodicity of the excitation signal u is high, and if gain b isless than 0.5, then periodicity is low.

Another efficient way to derive the filter F(z) coefficients used in apreferred embodiment, is to relate them to the amount of pitchcontribution in the total excitation signal u. This results in afrequency response depending on the subframe periodicity, where higherfrequencies are more strongly emphasized (stronger overall slope) forhigher pitch gains. Innovation filter 205 has the effect of lowering theenergy of the innovative codevector c_(k) at low frequencies when theexcitation signal u is more periodic, which enhances the periodicity ofthe excitation signal u at lower frequencies more than higherfrequencies. Suggested forms for innovation filter 205 areF(z)=1−σz ⁻¹,  (1)orF(z)=−αz+1−αz ⁻¹  (2)where σ or α are periodicity factors derived from the level ofperiodicity of the excitation signal u.

The second three-term form of F(z) is used in a preferred embodiment.The periodicity factor α is computed in the voicing factor generator204. Several methods can be used to derive the periodicity factor αbased on the periodicity of the excitation signal u. Two methods arepresented below.

Method 1:

The ratio of pitch contribution to the total excitation signal u isfirst computed in voicing factor generator 204 by$R_{p} = {\frac{b^{2}v_{T}^{t}v_{T}}{u^{t}u} = \frac{b^{2}{\sum\limits_{n = 0}^{N - 1}{v_{T}^{2}(n)}}}{\sum\limits_{n = 0}^{N - 1}{u^{2}(n)}}}$where V_(T) is the pitch codebook vector, b is the pitch gain, and u isthe excitation signal u given at the output of the adder 219 byu=gc _(k) +bv _(T)

Note that the term bv_(T) has its source in the pitch codebook (pitchcodebook) 201 in response to the pitch lag T and the past value of ustored in memory 203. The pitch codevector v_(T) from the pitch codebook201 is then processed through a low-pass filter 202 whose cut-offfrequency is adjusted by means of the index j from the demultiplexer217. The resulting codevector v_(T) is then multiplied by the gain bfrom the demultiplexer 217 through an amplifier 226 to obtain the signalbv_(T).

The factor α is calculated in voicing factor generator 204 byα=qR _(p) bounded by α<qwhere q is a factor which controls the amount of enhancement (q is setto 0.25 in this preferred embodiment).Method 2:

Another method used in a preferred embodiment of the invention forcalculating periodicity factor α is discussed below.

First, a voicing factor r_(v) is computed in voicing factor generator204 byr _(v)=(E _(v) −E _(c))/(E _(v) +E _(c))where E_(v) is the energy of the scaled pitch codevector bV_(T) andE_(c) is the energy of the scaled innovative codevector gc_(k). That is$E_{v} = {{b^{2}v_{T}^{t}v_{T}} = {{b^{2}{\sum\limits_{n = 0}^{N - 1}{{v_{T}^{2}(n)}\quad{and}\quad E_{c}}}} = {{g^{2}c_{k}^{t}c_{k}} = {g^{2}{\sum\limits_{n = 0}^{N - 1}{{c_{k}^{2}(n)}.}}}}}}$

Note that the value of r_(v) lies between −1 and 1 (1 corresponds topurely voiced signals and −1 corresponds to purely unvoiced signals).

In this preferred embodiment, the factor α is then computed in voicingfactor generator 204 byα=0.125 (1+r _(v))which corresponds to a value of 0 for purely unvoiced signals and 0.25for purely voiced signals.

In the first, two-term form of F(z), the periodicity factor σ can beapproximated by using σ=2α in methods 1 and 2 above. In such a case, theperiodicity factor σ is calculated as follows in method 1 above:σ=2qR _(p) bounded by σ<2q.

In method 2, the periodicity factor σ is calculated as follows:σ=0.25 (1+r _(v)).

The enhanced signal c_(f) is therefore computed by filtering the scaledinnovative codevector gc_(k) through the innovation filter 205 (F(z)).

The enhanced excitation signal u′ is computed by the adder 220 as:u′=c _(f) +bv _(T)

Note that this process is not performed at the encoder 100. Thus, it isessential to update the content of the pitch codebook 201 using theexcitation signal u without enhancement to keep synchronism between theencoder 100 and decoder 200. Therefore, the excitation signal u is usedto update the memory 203 of the pitch codebook 201 and the enhancedexcitation signal u′ is used at the input of the LP synthesis filter206.

Synthesis and Deemphasis

The synthesized signal s′ is computed by filtering the enhancedexcitation signal u′ through the LP synthesis filter 206 which has theform 1/Â(z), where Â(z) is the interpolated LP filter in the currentsubframe. As can be seen in FIG. 2, the quantized LP coefficients Â(z)on line 225 from demultiplexer 217 are supplied to the LP synthesisfilter 206 to adjust the parameters of the LP synthesis filter 206accordingly. The deemphasis filter 207 is the inverse of the preemphasisfilter 103 of FIG. 1. The transfer function of the deemphasis filter 207is given byD(z)=1/(1−μz ⁻¹)where μ is a preemphasis factor with a value located between 0 and 1 (atypical value is μ=0.7). A higher-order filter could also be used.

The vector s′ is filtered through the deemphasis filter D(z). (module207) to obtain the vector s_(d) which is passed through the high-passfilter 208 to remove the unwanted frequencies below 50 Hz and furtherobtain S_(h).

Oversampling and High-Frequency Regeneration

The over-sampling module 209 conducts the inverse process of thedown-sampling module 101 of FIG. 1. In this preferred embodiment,oversampling converts from the 12.8 kHz sampling rate to the original 16kHz sampling rate, using techniques well known to those of ordinaryskill in the art. The oversampled synthesis signal is denoted ŝ. Signalŝ is also referred to as the synthesized wideband intermediate signal.

The oversampled synthesis ŝ signal does not contain the higher frequencycomponents which were lost by the downsampling process (module 101 ofFIG. 1) at the encoder 100. This gives a low-pass perception to thesynthesized speech signal. To restore the full band of the originalsignal, a high frequency generation procedure is disclosed. Thisprocedure is performed in modules 210 to 216, and adder 221, andrequires input from voicing factor generator 204 (FIG. 2).

In this new approach, the high frequency contents are generated byfilling the upper-part of the spectrum with a white noise properlyscaled in the excitation domain, then converted to the speech domain,preferably by shaping it with the same LP synthesis filter used forsynthesizing the down-sampled signal ŝ.

The high frequency generation procedure in accordance with the presentinvention is described hereinbelow.

The random noise generator 213 generates a white noise sequence w′ witha flat spectrum over the entire frequency bandwidth, using techniqueswell known to those of ordinary skill in the art. The generated sequenceis of length N′ which is the subframe length in the original domain.Note that N is the subframe length in the down-sampled domain. In thispreferred embodiment, N=64 and N′=80 which correspond to 5 ms.

The white noise sequence is properly scaled in the gain adjusting module214. Gain adjustment comprises the following steps. First, the energy ofthe generated noise sequence w′ is set equal to the energy of theenhanced excitation signal u′ computed by an energy computing module210, and the resulting scaled noise sequence is given by${{w(n)} = {{w^{\prime}(n)}\sqrt{\frac{\sum\limits_{n = 0}^{N - 1}{u^{\prime 2}(n)}}{\sum\limits_{n = 0}^{N^{\prime} - 1}{w^{\prime 2}(n)}}}}},\quad{n = 0},\ldots\quad,{N^{\prime} - 1.}$

The second step in the gain scaling is to take into account the highfrequency contents of the synthesized signal at the output of thevoicing factor generator 204 so as to reduce the energy of the generatednoise in case of voiced segments (where less energy is present at highfrequencies compared to unvoiced segments). In this preferredembodiment, measuring the high frequency contents is implemented bymeasuring the tilt of the synthesis signal through a spectral tiltcalculator 212 and reducing the energy accordingly. Other measurementssuch as zero crossing measurements can equally be used. When the tilt isvery strong, which corresponds to voiced segments, the noise energy isfurther reduced. The tilt factor is computed in module 212 as the firstcorrelation coefficient of the synthesis signal s_(h) and it is givenby:${{tilt} = \frac{\sum\limits_{n = 1}^{N - 1}{{s_{h}(n)}{s_{h}\left( {n - 1} \right)}}}{\sum\limits_{n = 1}^{N - 1}{s_{h}^{2}(n)}}},{{{conditioned}\quad{by}\quad{tilt}} \geq {0\quad{and}\quad{tilt}} \geq {r_{v}.}}$where voicing factor r_(v) is given byr _(v)=(E _(v) −E _(c))/(E _(v) +E _(c))where E_(v) is the energy of the scaled pitch codevector bv_(T) andE_(c) is the energy of the scaled innovative codevector gc_(k), asdescribed earlier. Voicing factor r_(v) is most often less than tilt butthis condition was introduced as a precaution against high frequencytones where the tilt value is negative and the value of r_(v) is high.Therefore, this condition reduces the noise energy for such tonalsignals.

The tilt value is 0 in case of flat spectrum and 1 in case of stronglyvoiced signals, and it is negative in case of unvoiced signals wheremore energy is present at high frequencies.

Different methods can be used to derive the scaling factor g_(t) fromthe amount of high frequency contents. In this invention, two methodsare given based on the tilt of signal described above.

Method 1:

The scaling factor g_(t) is derived from the tilt byg _(t)=1−tilt bounded by 0.2≦g_(t)≦1.0For strongly voiced signal where the tilt approaches 1, g_(t) is 0.2 andfor strongly unvoiced signals g_(t) becomes 1.0.

Method 2:

The tilt factor g_(t) is first restricted to be larger or equal to zero,then the scaling factor is derived from the tilt byg_(t)=10^(−tilt)

The scaled noise sequence w_(g) produced in gain adjusting module 214 istherefore given by:w _(g) =g _(t) w.

When the tilt is close to zero, the scaling factor g_(t) is close to 1,which does not result in energy reduction. When the tilt value is 1, thescaling factor g_(t) results in a reduction of 12 dB in the energy ofthe generated noise.

Once the noise is properly scaled (w_(g)), it is brought into the speechdomain using the spectral shaper 215. In the preferred embodiment, thisis achieved by filtering the noise w_(g) through a bandwidth expandedversion of the same LP synthesis filter used in the down-sampled domain(1/Â(z/0.8)). The corresponding bandwidth expanded LP filtercoefficients are calculated in spectral shaper 215.

The filtered scaled noise sequence w_(f) is then band-pass filtered tothe required frequency range to be restored using the band-pass filter216. In the preferred embodiment, the band-pass filter 216 restricts thenoise sequence to the frequency range 5.6-7.2 kHz. The resultingband-pass filtered noise sequence z is added in adder 221 to theoversampled synthesized speech signal ŝ to obtain the finalreconstructed sound signal s_(out) on the output 223.

Although the present invention has been described hereinabove by way ofa preferred embodiment thereof, this embodiment can be modified at will,within the scope of the appended claims, without departing from thespirit and nature of the subject invention. Even though the preferredembodiment discusses the use of wideband speech signals, it will beobvious to those skilled in the art that the subject invention is alsodirected to other embodiments using wideband signals in general and thatit is not necessarily limited to speech applications.

1. A pitch analysis device for producing an optimal set of pitchcodebook parameters, comprising: a) at least two signal paths associatedto respective sets of pitch codebook parameters, wherein: i) each signalpath comprises a pitch prediction error calculating device forcalculating a pitch prediction error of a pitch codevector from a pitchcodebook search device; and ii) at least one of said two paths comprisesa filter for filtering the pitch codevector before supplying said pitchcodevector to the pitch prediction error calculating device of said onepath; and b) a selector for comparing the pitch prediction errorscalculated in said at least two signal paths, for choosing the signalpath having the lowest calculated pitch prediction error, and forselecting the set of pitch codebook parameters associated to the choosensignal path.
 2. A pitch analysis device as defined in claim 1, whereinone of said at least two paths comprises no filter for filtering thepitch codevector before supplying said pitch codevector to the pitchprediction error calculating device.
 3. A pitch analysis device asdefined in claim 1, wherein said signal paths comprises a plurality ofsignal paths each provided with a filter for filtering the pitchcodevector before supplying said pitch codevector to the pitchprediction error calculating device of the same path.
 4. A pitchanalysis device as defined in claim 3, wherein the filters of saidplurality of paths are selected from the group consisting of low-passand band-pass filters, and wherein said filters have different frequencyresponses.
 5. A pitch analysis device as defined in claim 1, whereineach pitch prediction error calculating device comprises: a) aconvolution unit for convolving the pitch codevector with a weightedsynthesis filter impulse response signal and therefore calculating aconvolved pitch codevector; b) a pitch gain calculator for calculating apitch gain in response to the convolved pitch codevector and a pitchsearch target vector; c) an amplifier for multiplying the convolvedpitch codevector by the pitch gain to thereby produce an amplifiedconvolved pitch codevector; and d) a combiner circuit for combining theamplified convolved pitch codevector with the pitch search target vectorto thereby produce the pitch prediction error.
 6. A pitch analysisdevice as defined in claim 5, wherein said pitch gain calculatorcomprises a means for calculating said pitch gain b^((j)) using therelation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector. 7.A pitch analysis device as defined in claim 1, wherein said pitchprediction error calculating device of each signal path comprises meansfor calculating an energy of the corresponding pitch prediction error,and wherein said selector comprises means for comparing the energies ofsaid pitch prediction errors of the different signal paths and forchoosing as the signal path having the lowest calculated pitchprediction error the signal path having the lowest calculated energy ofthe pitch prediction error.
 8. A pitch analysis device as defined inclaim 5, wherein: a) each of said filters of the plurality of signalpaths is identified by a filter index; b) said pitch codevector isidentified by a pitch codebook index; and c) said pitch codebookparameters comprise the filter index, the pitch codebook index and thepitch gain.
 9. A pitch analysis device as defined in claim 1, whereinsaid filter is integrated in an interpolation filter of said pitchcodebook search device, said interpolation filter being used to producea sub-sample version of said pitch codevector.
 10. A pitch analysismethod for producing an optimal set of pitch codebook parameters,comprising: a) in at least two signal paths associated to respectivesets of pitch codebook parameters, calculating, for each signal path, apitch prediction error of a pitch codevector from a pitch codebooksearch device; b) in at least one of said two signal paths, filteringthe pitch codevector before supplying said pitch codevector forcalculation of said pitch prediction error of said one path; and c)comparing the pitch prediction errors calculated in said at least twosignal paths, choosing the signal path having the lowest calculatedpitch prediction error, and selecting the set of pitch codebookparameters associated to the choosen signal path.
 11. A pitch analysismethod as defined in claim 10, wherein, in one of said at least twopaths, no filtering of the pitch codevector is performed beforesupplying said pitch codevector to the pitch prediction errorcalculating device.
 12. A pitch analysis method as defined in claim 10,wherein said signal paths comprises a plurality of signal paths andwherein filtering the pitch codevector is performed in each of saidplurality of signal paths before supplying said pitch codevector to thepitch prediction error calculating device of the same path.
 13. A pitchanalysis method as defined in claim 12, further comprising selecting thefilters of said plurality of paths from the group consisting of low-passand band-pass filters, and wherein said filters have different frequencyresponses.
 14. A pitch analysis method as defined in claim 10, whereincalculating a pitch prediction error in each signal path comprises: a)convolving the pitch codevector with a weighted synthesis filter impulseresponse signal and therefore calculating a convolved pitch codevector;b) calculating a pitch gain in response to the convolved pitchcodevector and a pitch search target vector; c) multiplying theconvolved pitch codevector by the pitch gain to thereby produce anamplified convolved pitch codevector; and d) combining the amplifiedconvolved pitch codevector with the pitch search target vector tothereby produce the pitch prediction error.
 15. A pitch analysis methodas defined in claim 14, wherein said pitch gain calculation comprisescalculating said pitch gain b^((j)) using the relation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.16. A pitch analysis method as defined in claim 10, wherein calculatingsaid pitch prediction error, in each signal path, comprises calculatingan energy of the corresponding pitch prediction error, and whereincomparing the pitch prediction error comprises comparing the energies ofsaid pitch prediction errors of the different signal paths and choosingas the signal path having the lowest calculated pitch prediction errorthe signal path having the lowest calculated energy of the pitchprediction error.
 17. A pitch analysis method as defined in claim 14,wherein: a) identifying each of said filters of the plurality of signalpaths by a filter index; b) identifying said pitch codevector by a pitchcodebook index; and c) said pitch codebook parameters comprise thefilter index, the pitch codebook index and the pitch gain.
 18. A pitchanalysis method as defined in claim 10, wherein said filtering the pitchcodevector is integrated in an interpolation filter of said pitchcodebook search device, said interpolation filter being used to producea sub-sample version of said pitch codevector.
 19. An encoder having apitch analysis device as in claim 1 for encoding a wideband inputsignal, said encoder comprising: a) a linear prediction synthesis filtercalculator responsive to the wideband signal for producing linearprediction synthesis filter coefficients; b) a perceptual weightingfilter, responsive to the wideband signal and the linear predictionsynthesis filter coefficients, for producing a perceptually weightedsignal; c) an impulse response generator responsive to said linearprediction synthesis filter coefficients for producing a weightedsynthesis filter impulse response signal; d) a pitch search unit forproducing pitch codebook parameters, said pitch search unit comprising:i) said pitch codebook search device responsive to the perceptuallyweighted signal and the linear prediction synthesis filter coefficientsfor producing the pitch codevector and an innovative search targetvector; and ii) said pitch analysis device responsive to the pitchcodevector for selecting, from said sets of pitch codebook parameters,the set of pitch codebook parameters associated to the path having thelowest calculated pitch prediction error; d) an innovative codebooksearch device, responsive to the weighted synthesis filter impulseresponse signal, and the innovative search target vector, for producinginnovative codebook parameters; and e) a signal forming device forproducing an encoded wideband signal comprising the set of pitchcodebook parameters associated to the path having the lowest pitchprediction error, said innovative codebook parameters, and said linearprediction synthesis filter coefficients.
 20. An encoder as defined inclaim 19, wherein one of said at least two paths comprises no filter forfiltering the pitch codevector before supplying said pitch codevector tothe pitch prediction error calculating device.
 21. An encoder as definedin claim 19, wherein said signal paths comprises a plurality of signalpaths each provided with a filter for filtering the pitch codevectorbefore supplying said pitch codevector to the pitch prediction errorcalculating device of the same path.
 22. An encoder as defined in claim21, wherein the filters of said plurality of paths are selected from thegroup consisting of low-pass and band-pass filters, and wherein saidfilters have different frequency responses.
 23. An encoder as defined inclaim 19, wherein each pitch prediction error calculating devicecomprises: a) a convolution unit for convolving the pitch codevectorwith the weighted synthesis filter impulse response signal and thereforecalculating a convolved pitch codevector; b) a pitch gain calculator forcalculating a pitch gain in response to the convolved pitch codevectorand the pitch search target vector; c) an amplifier for multiplying theconvolved pitch codevector by the pitch gain to thereby produce anamplified convolved pitch codevector; and d) a combiner circuit forcombining the amplified convolved pitch codevector with the pitch searchtarget vector to thereby produce the pitch prediction error.
 24. Anencoder as defined in claim 23, wherein said pitch gain calculatorcomprises a means for calculating said pitch gain b^((j)) using therelation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.25. An encoder as defined in claim 19, wherein said pitch predictionerror calculating device of each signal path comprises means forcalculating an energy of the corresponding pitch prediction error, andwherein said selector comprises means for comparing the energies of saidpitch prediction errors of the different signal paths and for choosingas the signal path having the lowest calculated pitch prediction errorthe signal path having the lowest calculated energy of the pitchprediction error.
 26. An encoder as defined in claim 23, wherein: a)each of said filters of the plurality of signal paths is identified by afilter index; b) said pitch codevector is identified by a pitch codebookindex; and c) said pitch codebook parameters comprise the filter index,the pitch codebook index and the pitch gain.
 27. A encoder as defined inclaim 19, wherein said filter is integrated in an interpolation filterof said pitch codebook search device, said interpolation filter beingused to produce a sub-sample version of said pitch codevector.
 28. Acellular communication system for servicing a large geographical areadivided into a plurality of cells, comprising: a) mobiletransmitter/receiver units; b) cellular base stations respectivelysituated in said cells; c) a control terminal for controllingcommunication between the cellular base stations; d) a bidirectionalwireless communication sub-system between each mobile unit situated inone cell and the cellular base station of said one cell, saidbidirectional wireless communication sub-system comprising, in both themobile unit and the cellular base station: i) a transmitter including anencoder for encoding a wideband signal as recited in claim 19 and atransmission circuit for transmitting the encoded wideband signal; andii) a receiver including a receiving circuit for receiving a transmittedencoded wideband signal and a decoder for decoding the received encodedwideband signal.
 29. A cellular communication system as defined in claim28, wherein one of said at least two paths comprises no filter forfiltering the pitch codevector before supplying said pitch codevector tothe pitch prediction error calculating device.
 30. A cellularcommunication system as defined in claim 28, wherein said signal pathscomprises a plurality of signal paths each provided with a filter forfiltering the pitch codevector before supplying said pitch codevector tothe pitch prediction error calculating device of the same path.
 31. Acellular communication system as defined in claim 30, wherein thefilters of said plurality of paths are selected from the groupconsisting of low-pass and band-pass filters, and wherein said filtershave different frequency responses.
 32. A cellular communication systemas defined in claim 28, wherein each pitch prediction error calculatingdevice comprises: a) a convolution unit for convolving the pitchcodevector with the weighted synthesis filter impulse response signaland therefore calculating a convolved pitch codevector; b) a pitch gaincalculator for calculating a pitch gain in response to the convolvedpitch codevector and the pitch search target vector; c) an amplifier formultiplying the convolved pitch codevector by the pitch gain to therebyproduce an amplified convolved pitch codevector; and d) a combinercircuit for combining the amplified convolved pitch codevector with thepitch search target vector to thereby produce the pitch predictionerror.
 33. A cellular communication system as defined in claim 32,wherein said pitch gain calculator comprises a means for calculatingsaid pitch gain b^((j)) using the relation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.34. A cellular communication system as defined in claim 28, wherein saidpitch prediction error calculating device of each signal path comprisesmeans for calculating an energy of the corresponding pitch predictionerror, and wherein said selector comprises means for comparing theenergies of said pitch prediction errors of the different signal pathsand for choosing as the signal path having the lowest calculated pitchprediction error the signal path having the lowest calculated energy ofthe pitch prediction error.
 35. A cellular communication system asdefined in claim 32, wherein: a) each of said filters of the pluralityof signal paths is identified by a filter index; b) said pitchcodevector is identified by a pitch codebook index; and c) said pitchcodebook parameters comprise the filter index, the pitch codebook indexand the pitch gain.
 36. A cellular communication system as defined inclaim 28, wherein said filter is integrated in an interpolation filterof said pitch codebook search device, said interpolation filter beingused to produce a sub-sample version of said pitch codevector.
 37. Acellular mobile transmitter/receiver unit comprising: a) a transmitterincluding an encoder for encoding a wideband signal as recited in claim19 and a transmission circuit for transmitting the encoded widebandsignal; and b) a receiver including a receiving circuit for receiving atransmitted encoded wideband signal and a decoder for decoding thereceived encoded wideband signal.
 38. A cellular mobiletransmitter/receiver unit as defined in claim 37, wherein one of said atleast two paths comprises no filter for filtering the pitch codevectorbefore supplying said pitch codevector to the pitch prediction errorcalculating device.
 39. A cellular mobile transmitter/receiver unit asdefined in claim 37, wherein said signal paths comprises a plurality ofsignal paths each provided with a filter for filtering the pitchcodevector before supplying said pitch codevector to the pitchprediction error calculating device of the same path.
 40. A cellularmobile transmitter/receiver unit as defined in claim 39, wherein thefilters of said plurality of paths are selected from the groupconsisting of low-pass and band-pass filters, and wherein said filtershave different frequency responses.
 41. A cellular mobiletransmitter/receiver unit as defined in claim 37, wherein each pitchprediction error calculating device comprises: a) a convolution unit forconvolving the pitch codevector with the weighted synthesis filterimpulse response signal and therefore calculating a convolved pitchcodevector; b) a pitch gain calculator for calculating a pitch gain inresponse to the convolved pitch codevector and the pitch search targetvector; c) an amplifier for multiplying the convolved pitch codevectorby the pitch gain to thereby produce an amplified convolved pitchcodevector; and d) a combiner circuit for combining the amplifiedconvolved pitch codevector with the pitch search target vector tothereby produce the pitch prediction error.
 42. A cellular mobiletransmitter/receiver unit as defined in claim 41, wherein said pitchgain calculator comprises a means for calculating said pitch gainb^((j)) using the relation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.43. A cellular mobile transmitter/receiver unit as defined in claim 37,wherein said pitch prediction error calculating device of each signalpath comprises means for calculating an energy of the correspondingpitch prediction error, and wherein said selector comprises means forcomparing the energies of said pitch prediction errors of the differentsignal paths and for choosing as the signal path having the lowestcalculated pitch prediction error the signal path having the lowestcalculated energy of the pitch prediction error.
 44. A cellular mobiletransmitter/receiver unit as defined in claim 41, wherein: a) each ofsaid filters of the plurality of signal paths is identified by a filterindex; b) said pitch codevector is identified by a pitch codebook index;and c) said pitch codebook parameters comprise the filter index, thepitch codebook index and the pitch gain.
 45. A cellular mobiletransmitter/receiver unit as defined in claim 37, wherein said filter isintegrated in an interpolation filter of said pitch codebook searchdevice, said interpolation filter being used to produce a sub-sampleversion of said pitch codevector.
 46. A cellular network elementcomprising: a) a transmitter including an encoder for encoding awideband signal as recited in claim 19 and a transmission circuit fortransmitting the encoded wideband signal; and b) a receiver including areceiving circuit for receiving a transmitted encoded wideband signaland a decoder for decoding the received encoded wideband signal.
 47. Acellular network element as defined in claim 46, wherein one of said atleast two paths comprises no filter for filtering the pitch codevectorbefore supplying said pitch codevector to the pitch prediction errorcalculating device.
 48. A cellular network element as defined in claim46, wherein said signal paths comprises a plurality of signal paths eachprovided with a filter for filtering the pitch codevector beforesupplying said pitch codevector to the pitch prediction errorcalculating device of the same path.
 49. A cellular network element asdefined in claim 48, wherein the filters of said plurality of paths areselected from the group consisting of low-pass and band-pass filters,and wherein said filters have different frequency responses.
 50. Acellular network element as defined in claim 46, wherein each pitchprediction error calculating device comprises: a) a convolution unit forconvolving the pitch codevector with the weighted synthesis filterimpulse response signal and therefore calculating a convolved pitchcodevector; b) a pitch gain calculator for calculating a pitch gain inresponse to the convolved pitch codevector and the pitch search targetvector; c) an amplifier for multiplying the convolved pitch codevectorby the pitch gain to thereby produce an amplified convolved pitchcodevector; and d) a combiner circuit for combining the amplifiedconvolved pitch codevector with the pitch search target vector tothereby produce the pitch prediction error.
 51. A cellular networkelement as defined in claim 50, wherein said pitch gain calculatorcomprises a means for calculating said pitch gain b^((j)) using therelation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.52. A cellular network element as defined in claim 46, wherein saidpitch prediction error calculating device of each signal path comprisesmeans for calculating an energy of the corresponding pitch predictionerror, and wherein said selector comprises means for comparing theenergies of said pitch prediction errors of the different signal pathsand for choosing as the signal path having the lowest calculated pitchprediction error the signal path having the lowest calculated energy ofthe pitch prediction error.
 53. A cellular network element as defined inclaim 50, wherein: a) each of said filters of the plurality of signalpaths is identified by a filter index; b) said pitch codevector isidentified by a pitch codebook index; and c) said pitch codebookparameters comprise the filter index, the pitch codebook index and thepitch gain.
 54. A cellular network element as defined in claim 46,wherein said filter is integrated in an interpolation filter of saidpitch codebook search device, said interpolation filter being used toproduce a sub-sample version of said pitch codevector.
 55. In a cellularcommunication system for servicing a large geographical area dividedinto a plurality of cells, comprising: mobile transmitter/receiverunits; cellular base stations, respectively situated in said cells; andcontrol terminal for controlling communication between the cellular basestations: a bidirectional wireless communication sub-system between eachmobile unit situated in one cell and the cellular base station of saidone cell, said bidirectional wireless communication sub-systemcomprising, in both the mobile unit and the cellular base station: a) atransmitter including an encoder for encoding a wideband signal asrecited in claim 19 and a transmission circuit for transmitting theencoded wideband signal; and b) a receiver including a receiving circuitfor receiving a transmitted encoded wideband signal and a decoder fordecoding the received encoded wideband signal.
 56. A bidirectionalwireless communication sub-system as defined in claim 55, wherein one ofsaid at least two paths comprises no filter for filtering the pitchcodevector before supplying said pitch codevector to the pitchprediction error calculating device.
 57. A bidirectional wirelesscommunication sub-system as defined in claim 55, wherein said signalpaths comprises a plurality of signal paths each provided with a filterfor filtering the pitch codevector before supplying said pitchcodevector to the pitch prediction error calculating device of the samepath.
 58. A bidirectional wireless communication sub-system as definedin claim 57, wherein the filters of said plurality of paths are selectedfrom the group consisting of low-pass and band-pass filters, and whereinsaid filters have different frequency responses.
 59. A bidirectionalwireless communication sub-system as defined in claim 55, wherein eachpitch prediction error calculating device comprises: a) a convolutionunit for convolving the pitch codevector with the weighted synthesisfilter impulse response signal and therefore calculating a convolvedpitch codevector; b) a pitch gain calculator for calculating a pitchgain in response to the convolved pitch codevector and the pitch searchtarget vector; c) an amplifier for multiplying the convolved pitchcodevector by the pitch gain to thereby produce an amplified convolvedpitch codevector; and d) a combiner circuit for combining the amplifiedconvolved pitch codevector with the pitch search target vector tothereby produce the pitch prediction error.
 60. A bidirectional wirelesscommunication sub-system as defined in claim 59, wherein said pitch gaincalculator comprises a means for calculating said pitch gain b^((j))using the relation:b ^((j)) =x ^(t) y ^((j)) /∥y ^((j))∥² where j=0, 1, 2, . . . , K, and Kcorresponds to a number of signal paths, and where x is said pitchsearch target vector, and y^((j)) is said convolved pitch codevector.61. A bidirectional wireless communication sub-system as defined inclaim 55, wherein said pitch prediction error calculating device of eachsignal path comprises means for calculating an energy of thecorresponding pitch prediction error, and wherein said selectorcomprises means for comparing the energies of said pitch predictionerrors of the different signal paths and for choosing as the signal pathhaving the lowest calculated pitch prediction error the signal pathhaving the lowest calculated energy of the pitch prediction error.
 62. Abidirectional wireless communication sub-system as defined in claim 59,wherein: a) each of said filters of the plurality of signal paths isidentified by a filter index; b) said pitch codevector is identified bya pitch codebook index; and c) said pitch codebook parameters comprisethe filter index, the pitch codebook index and the pitch gain.
 63. Abidirectional wireless communication sub-system as defined in claim 55,wherein said filter is integrated in an interpolation filter of saidpitch codebook search device, said interpolation filter being used toproduce a sub-sample version of said pitch codevector.